LM27341_15 datasheet(14/44 Pages) TI

14 / 44 page

r =

'iL

lOUT

LM27341, LM27342, LM27341-Q1, LM27342-Q1

SNVS497E – NOVEMBER 2008 – REVISED APRIL 2013

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OUTPUT OVERVOLTAGE PROTECTION

The overvoltage comparator turns off the internal power NFET when the FB pin voltage exceeds the internal

reference voltage by 13% (VFB > 1.13 * VREF). With the power NFET turned off the output voltage will decrease

toward the regulation level.

UNDERVOLTAGE LOCKOUT

Undervoltage lockout (UVLO) prevents the LM27341/LM27342 from operating until the input voltage exceeds

2.75V(typ).

The UVLO threshold has approximately 470 mV of hysteresis, so the part will operate until VIN drops below

2.28V(typ). Hysteresis prevents the part from turning off during power up if VIN has finite impedance.

THERMAL SHUTDOWN

Thermal shutdown limits total power dissipation by turning off the internal NMOS switch when the IC junction

temperature exceeds 165°C (typ). After thermal shutdown occurs, hysteresis prevents the internal NMOS switch

from turning on until the junction temperature drops to approximately 150°C.

Design Guide

INDUCTOR SELECTION

Inductor selection is critical to the performance of the LM27341/LM27342. The selection of the inductor affects

stability, transient response and efficiency. A key factor in inductor selection is determining the ripple current (

ΔiL)

(see Figure 32).

The ripple current (

ΔiL) is important in many ways.

First, by allowing more ripple current, lower inductance values can be used with a corresponding decrease in

physical dimensions and improved transient response. On the other hand, allowing less ripple current will

increase the maximum achievable load current and reduce the output voltage ripple (see OUTPUT CAPACITOR

section for more details on calculating output voltage ripple). Increasing the maximum load current is achieved by

ensuring that the peak inductor current (ILPK) never exceeds the minimum current limit of 2.0A min (LM27341) or

2.5A min (LM27342) .

ILPK = IOUT + ΔiL / 2

(8)

Secondly, the slope of the ripple current affects the current control loop. The LM27341/LM27342 has a fixed

slope corrective ramp. When the slope of the current ripple becomes significantly less than the converter’s

corrective ramp (see Figure 31), the inductor pole will move from high frequencies to lower frequencies. This

negates one advantage that peak current-mode control has over voltage-mode control, which is, a single low

frequency pole in the power stage of the converter. This can reduce the phase margin, crossover frequency and

potentially cause instability in the converter. Contrarily, when the slope of the ripple current becomes significantly

greater than the converter’s corrective ramp, resonant peaking can occur in the control loop. This can also cause

instability (Sub-Harmonic Oscillation) in the converter. For the power supply designer this means that for lower

switching frequencies the current ripple must be increased to keep the inductor pole well above crossover. It also

means that for higher switching frequencies the current ripple must be decreased to avoid resonant peaking.

With all these factors, how is the desired ripple current selected? The ripple ratio (r) is defined as the ratio of

inductor ripple current (

ΔiL) to output current (IOUT), evaluated at maximum load:

(9)

A good compromise between physical size, transient response and efficiency is achieved when we set the ripple

ratio between 0.2 and 0.4. The recommended ripple ratio vs. duty cycle shown below (see Figure 36) is based

upon this compromise and control loop optimizations. Note that this is just a guideline. Please see Application

note AN-1197 SNVA038 for further considerations.

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